In a wireless communication system for the next-generation mobile communications, it is important to achieve high-speed data transmission. As a technology to realize high-speed data transmission, the MIMO (Multiple Input Multiple Output) multiplexing system attracts attention. According to the MIMO multiplexing system, signals are transmitted at the same frequency and time through a plurality of transmit antennas, and the signals are demodulated (separated) through a plurality of receive antennas.
FIG. 1 is a diagram showing a MIMO transmitter/receiver having M (M: an integer not less than 1) transmit antennas and N (N: an integer not less than 1) receive antennas. Referring to FIG. 1, the MIMO transmitter/receiver comprises transmit antennas 1-1 to 1-M and a transmitter 2 on the transmitting side, and receive antennas 3-1 to 3-N and a receiver 4 on the receiving side. The MIMO transmitter/receiver transmits different signals through the plurality of transmit antennas 1-1 to 1-M at the same frequency and time, and receives them through the plurality of receive antennas 3-1 to 3-N. Thereby, the MIMO transmitter/receiver is capable of high-speed data transmission in proportion to the number of transmit antennas without any increase in transmission bandwidth. The receiving side is required to perform signal separation to demodulate the signals received through the receive antennas 3-1 to 3-N to the original signals from the transmit antennas 1-1 to 1-M.
There are a variety of methods to demodulate a MIMO multiplexed signal. Among them is linear filter reception, which can be relatively easily conducted. When the MIMO multiplexing is applied to DS-CDMA (Direct Sequence-Code Division Multiple Access) signals, in addition to interference from other transmit antennas, the multipath of desired transmit antenna signals interferes with the DS-CDMA signals. The filter reception is advantageous to suppress such interference all at once. There has been proposed a frequency-domain equalizer that performs the signal processing in the frequency domain by a simple method. Reference may be had to, for example, Xu Zhu and Ross D. Murch, “Novel Frequency-Domain Equalization Architectures for a Single-Carrier Wireless MIMO System,” IEEE VTC2002-Fall, pp. 874-878, September 2002.
FIG. 2 is a diagram showing an example of the construction of a DS-CDMA MIMO receiver to which is applied the aforementioned frequency-domain equalizer. In the following, a description will be given of the MIMO receiver with M (M: an integer not less than 1) transmit antennas and N (N: an integer not less than 1) receive antennas. As can be seen in FIG. 2, the conventional MIMO receiver comprises transmission channel estimation sections 10-1-1 to 10-M-N, S/P (Serial to Parallel) converters 11-1-1 to 11-M-N and 14-1 to 14-N, FFT (Fast Fourier Transform) sections 12-1-1 to 12-M-N and 15-1 to 15-N, GI (Guard Interval) removal sections 13-1 to 13-N, a chip noise estimation section 16, a weight calculator 17, a filter 18, IFFT (Inverse Fast Fourier Transform) sections 19-1 to 19-M, P/S (Parallel to Serial) converters 20-1 to 20-M, and despread circuits 21-1 to 21-M.
The transmission channel estimation sections 10-1-1 to 10-M-N input therein signals received by the receive antennas 3-1 to 3-N, respectively. With a known pilot signal contained in the received signal, the transmission channel estimation sections 10-1-1 to 10-M-N each estimate a transmission channel estimation value with respect to each path between the transmit antennas 1-1 to 1-M and the receive antennas 3-1 to 3-N to obtain the impulse response.
The S/P converters 11-1-1 to 11-M-N convert the impulse responses of the transmission channels estimated by the transmission channel estimation sections 10-1-1 to 10-M-N from serial to parallel.
The FFT sections 12-1-1 to 12-M-N receive as input the transmission channel impulse responses converted by the S/P converters 11-1-1 to 11-M-N, respectively, to transfer them into frequency domain.
The chip noise estimation section 16 inputs therein the signals received by the receive antennas 3-1 to 3-N as well as the transmission channel estimation values obtained by the transmission channel estimation sections 10-1-1 to 10-M-N to estimate chip noise power.
The weight calculator 17 receives as input the transmission channel impulse responses transferred into frequency domain by the FFT sections 12-1-1 to 12-M-N and the chip noise power estimated by the chip noise estimation section 16. The weight calculator 17 calculates filter weights according to the minimum mean square error (MMSE) criterion.
The GI removal sections 13-1 to 13-N input therein signals received by the receive antennas 3-1 to 3-N, and eliminate part of the received signals corresponding to GI based on receive path timing.
The S/P converters 14-1 to 14-N convert the received signals, from which GI has been removed by the GI removal sections 13-1 to 13-N, from serial to parallel.
The FFT sections 15-1 to 15-N receive as input the received signals converted by the S/P converters 14-1 to 14-N, respectively, to transfer them into frequency domain.
The filter 18 inputs therein the weights obtained by the weight calculator 17 and the received signals transferred into frequency domain by the FFT sections 15-1 to 15-N. The filter 18 performs filtering (equalization) of the received signals in frequency domain.
The IFFT sections 19-1 to 19-M receive as input the frequency-domain signals equalized by the filter 18 to transfer them back to time domain.
The P/S converters 20-1 to 20-M convert back the signals transferred into time domain from parallel to serial.
The despread circuits 21-1 to 21-M receive as input the time-domain signals converted by the P/S converters 20-1 to 20-M, and despread the signals to demodulate them to the original signals transmitted from the transmit antennas 1-1 to 1-M.
FIG. 3 is a block diagram showing the construction of the weight calculator 17 for a subcarrier f (1≦f≦F) after the FFT. The conventional weight calculator 17 for the subcarrier f comprises correlation matrix generators 30-1 to 30-M, a correlation matrix adder 31, a noise adder 32, an inverse matrix calculator 33, and weight generators 34-1 to 34-M. The weight calculator 17 for each subcarrier has the same construction as described above.
The correlation matrix generators 30-1 to 30-M receive as input the transmission channel estimation values between the transmit and receive antennas transferred into frequency domain by the FFT sections 12-1-1 to 12-M-N shown in FIG. 2. Each of the correlation matrix generators 30-1 to 30-M generates a correlation matrix from transmit to receive antennas with respect to each transmit antenna.
The correlation matrix adder 31 receives as input the correlation matrices for the respective transmit antennas 1-1 to 1-M generated by the correlation matrix generators 30-1 to 30-M to add up them.
The noise adder 32 receives as input the summation correlation matrix obtained by the correlation matrix adder 31, the chip noise power estimated by the chip noise estimation section 16 shown in FIG. 2, and chip power ratio. The noise adder 32 multiplies the chip noise power by the inverse of the chip power ratio, and adds the product to the correlation matrix.
The inverse matrix calculator 33 receives as input the correlation matrix to which a noise component has been added by the noise adder 32, and performs inverse matrix calculation.
The weight generators 34-1 to 34-M receive as input the inverse matrix obtained by the inverse matrix calculator 33 and transmission channel estimation values between the transmit and receive antennas transferred into frequency domain. Thereby, the weight generators 34-1 to 34-M generate filter weights.
Next, a description will be given in detail of the above processing through the use of mathematical expressions. The transmission channel vector Hm(f) between the transmit antenna m (1≦m≦M) and the receive antennas of the subcarrier f, in which the transmission channel impulse responses have been transferred into frequency domain by the FFT sections 12-1-1 to 12-M-N, is defined as follows:Hm(f)=[hm, 1(f),hm, 2(f), . . . , hm, N(f)]T  (1)where T denotes transpose. Besides, the received signal vector X(f) of the subcarrier f, in which the received signals have been transferred into frequency domain by the FFT sections 15-1 to 15-N, is defined as follows:X(f)=[x1(f),x2(f), . . . , xN(f)]T  (2)
The weight vector Wm(f) of the filter of the transmit antenna m for the subcarrier f calculated by the weight calculator 17 is given by the following expression:
                                          W            m                    ⁡                      (            f            )                          =                                            [                                                                    ∑                                                                  m                        ′                                            =                      1                                        M                                    ⁢                                                                                    H                                                  m                          ′                                                                    ⁡                                              (                        f                        )                                                              ⁢                                                                                  ⁢                                                                  H                                                  m                          ′                                                H                                            ⁡                                              (                        f                        )                                                                                            +                                                                            PN                      0                                                              P                      +                      D                                                        ⁢                  I                                            ]                                      -              1                                ⁢                                    H              m                        ⁡                          (              f              )                                                          (        3        )            where H denotes complex conjugate transpose, P is pilot power, D is data power, N0 is chip noise power, and I is identity or unit matrix.
The transmitted signal vector Y(f) in the subcarrier f of the signal which has undergone equalization and filtering at the filter 18 is expressed as follows:Y(f)=WH(f)×(f)  (4)where W(f) and Y(f) are defined as follows:W(f)=[W1(f),W2(f), . . . , WM(f)]T  (5)Y(f)=[Y1(f),Y2(f), . . . , YM(f)]T  (6)
The conventional MIMO receiver, however, has the following problem. In the calculation of filter weights by the weight calculator 17, the correlation matrix generators 30-1 to 30-M generate a correlation matrix with respect to each transmit antenna. When the correlation matrix adder 31 adds up the correlation matrices for all the transmit antennas, the values of them determine the degree of suppression of interference between the transmit antennas. The aforementioned frequency-domain equalizer calculates filter weights on the assumption that the respective transmit antennas use the same total chip signal power to pilot signal power ratio (chip power ratio) r=(P+D)/P. Consequently, when the transmit antennas use different chip power ratios (perform adaptive modulation, etc.), respectively, the MMSE weight precision decreases, and the characteristics deteriorate.